1. Field of the Invention
The present invention relates to a high-efficiency current-resonant type switching power supply.
2. Description of the Related Art
FIG. 1 is a circuit block diagram of a conventional current-resonant type switching power supply. In FIG. 1, a series circuit including switching elements QH and QL typically implemented by MOSFETs is connected to both ends of a direct-current power source 1. One end of the switching element QH is connected to a positive electrode of the direct-current power source 1. One end of the switching element QL is connected to a negative electrode of the direct-current power source 1.
A diode D1 is connected in parallel with the switching element QH. A diode D2 is connected in parallel with the switching element QL. Moreover, a voltage-resonant capacitor Crv is connected in parallel with the switching element QH.
Meanwhile, a current-resonant circuit formed of a current-resonant capacitor Cri, a resonant reactor Lr, and a primary winding Lp of a transformer T is connected in parallel with the voltage-resonant capacitor Crv. The voltage-resonant capacitor Crv, the current-resonant capacitor Cri, the resonant reactor Lr, and the primary winding Lp of the transformer T collectively constitute a resonant circuit.
Here, the resonant reactor Lr may be equivalent to leakage inductance of the transformer T. The voltage-resonant capacitor Crv may be equivalent to parasitic capacitance in the switching element QH. The diodes D1 and D2 connected in parallel with the switching elements may be equivalent to parasitic diodes in the respective switching elements.
The primary winding Lp and a secondary winding Ls are wound so as to generate common-mode voltages. A rectifying-smoothing circuit formed of a diode RC and a smoothing capacitor Co is connected to the secondary winding Ls of the transformer T. The rectifying-smoothing circuit rectifies and smoothes a voltage (an on/off controlled pulse voltage) induced in the secondary winding Ls of the transformer T, and outputs a direct-current output to a load 4.
An output voltage detection circuit 5 is connected to both ends of the smoothing capacitor Co. The output voltage detection circuit 5 detects an output voltage on the smoothing capacitor Co, and outputs an error voltage signal representing a difference between the detected voltage and a reference voltage to a pulse width modulation (PWM) control circuit 2 through a photocoupler PC. The PWM control circuit 2 regulates the voltage on the load 4 to a constant value by generating a PWM signal based on the error voltage signal from the output voltage detection circuit 5 and turning the switching elements QH and QL on and off alternately through a drive circuit 3 operated in accordance with the PWM signal. In this case, the PWM control circuit 2 turns the switching elements QH and QL on and off alternately by applying voltages to gates of the switching elements QH and QL alternately.
Next, operations of the conventional current-resonant type switching power supply will be described with reference to timing charts of FIGS. 2 and 3.
FIG. 2 shows timing charts of signals on respective portions before an input voltage to the conventional switching power supply is reduced. FIG. 3 shows timing charts of signals on the respective portions after the input voltage to the conventional switching power supply is reduced.
Here, in FIGS. 2 and 3, reference code ILp denotes a current flowing on the primary winding Lp. Reference code VQL denotes a voltage at both ends of the switching element QL. Reference code IQL denotes a current flowing on the switching element QL. Reference code IRC denotes a current flowing on the diode RC. Moreover, the resonant reactor Lr is sufficiently smaller than exciting inductance of the primary winding Lr, and the voltage-resonant capacitor Crv is sufficiently smaller than the current-resonant capacitor Cri.
First, when the switching element QL is turned on in a time frame T1, a current flows on a path in the order of the positive electrode of the direct-current power source 1, the current-resonant capacitor Cri, the primary winding Lp, the resonant reactor Lr, the switching element QL, and the negative electrode of the direct-current power source 1. At this time, the current IRC flows from the secondary winding Ls to the diode RC, and the voltage on the secondary winding Ls is rectified. The voltage rectified by the diode RC is smoothed by the capacitor Co, and the direct-current output is supplied to the load 4. Therefore, the current ILp which is equivalent to superimposition of a resonant current attributable to the current-resonant capacitor Cri and the resonant reactor Lr on an exciting current attributable to resonance among the primary winding Lp, the resonant reactor Lr, and the current-resonant capacitor Cri flows on the primary winding Lp of the transformer T (the same applies to the current IQL).
Next, the switching element QL remains turned on to charge the capacitor Co in a time frame T2. The diode RC is turned off when the current IRC stops flowing. A resonant current attributable to the current-resonant capacitor Cri, the exciting inductance of the primary winding Lp of the transformer T, and the resonant reactor Lr flows on the primary winding Lp of the transformer T as the current ILp in the form of a sinusoidal wave (the same applies to the current IQL).
Then, the switching element QL is turned off and the switching element QH is turned on in a time frame T3. At this time, charges accumulated in the exciting inductance of the primary winding Lp of the transformer T, the resonant reactor Lr, and the current-resonant capacitor Cri are discharged by the switching element QH as the resonant current attributable to the current-resonant capacitor Cri, the exciting inductance of the primary winding Lp of the transformer T, and the resonant reactor Lr. Then, the current ILp in the form of the sinusoidal wave flows and a core of the transformer T is reset.
Next, when the input voltage is set low, a boost rate is set high by expanding on-time of the switching element QH. However, the PWM control circuit 2 is performing PWM control. Therefore, as shown in FIG. 3, on-time of the switching element QL becomes shorter than that of the switching element QL shown in FIG. 2 as equivalent to the expanded amount of the on-time of the switching element QH. Accordingly, time for allowing only the resonant current attributable to the current-resonant capacitor Cri, the primary winding Lp of the transformer T, and the exciting inductance of the primary winding Lp of the transformer T to flow (corresponding to the time frame T2) becomes shorter.
When the input voltage is set even lower, time does not allow only the resonant current attributable to the current-resonant capacitor Cri, the primary winding Lp of the transformer T, and the exciting inductance of the primary winding Lp of the transformer T to flow. Therefore, the resonant current attributable to the current-resonant capacitor Cri and the resonant reactor Lr flows on the primary winding Lp of the transformer T, whereby the switching element QL on a primary side is turned off when energy is transmitted from the primary side to a secondary side of the transformer T. At this time, a current variation steeper than the resonant current attributable to the current-resonant capacitor Cri and the resonant reactor Lr is generated.
Meanwhile, U.S. Pat. No. 5,808,879 discloses a DC-DC converter. The DC-DC converter includes a half bridge of a semiconductor switch, and is configured to operate a voltage converter formed by connecting a series circuit including a primary winding of a transformer and a capacitor to the half bridge in accordance with PWM control.
Meanwhile, Japanese Unexamined Patent Publication No. 2003-9525 discloses another voltage converter. This voltage converter performs PVVM control by setting a capacity of a capacitor such that a resonance frequency attributable to the capacitor and leakage inductance of a transformer connected in series with a primary winding of the transformer becomes larger than a half value of an operating frequency of a semiconductor switch.
The DC-DC converter disclosed in U.S. Pat. No. 5,808,879 and the voltage converter disclosed in Japanese Unexamined Patent Publication No. 2003-9525 are configured to perform the PWM control by switching on-off states of the switching element on the primary side in a time period when a current is flowing on a diode RC on the secondary side. Accordingly, a current variation steeper than a resonant current attributable to a current-resonant capacitor Cri and a resonant reactor Lr is generated.
As described above, when the resonant type switching power supply configured to connect the semiconductor switch to the half bridge and simultaneously to connect the series circuit including the primary winding of the transformer and the capacitor to the half bridge as shown in FIG. 1 performs the PWM control, a timing for switching the switching element on the primary side takes place in a time period when a diode on the secondary side is conducted. Moreover, the current on the primary winding of the transformer (the current flowing on the switching element on the primary side) and a diode current sharply change as compared to the variation in the resonant current, which result in noise generation.
Moreover, the DC-DC converter disclosed in U.S. Pat. No. 5,808,879 and the voltage converter disclosed in Japanese Unexamined Patent Publication No. 2003-9525 are configured to switch the switching element on the primary side in a time period when energy is supplied from the primary side to the secondary side and when the current is flowing on the diode on the secondary side by the resonant current attributable to the resonant reactor and the current-resonant capacitor. At this time, the resonant current attributable to the exciting inductance of the primary winding of the transformer and the current-resonant capacitor as well as the resonant current attributable to the leakage inductance and the current-resonant capacitor are flowing on the switching element on the primary side. Therefore, large variations occur in the current on the primary winding of the transformer (the current flowing on the switching element on the primary side) and the diode current, which result in noise generation.